Power converters that convert three-phase AC input power to one or more DC power levels are widely known. The prior art power converters include high-power boost converters that output DC voltage levels that are higher than the peak of the AC input supply voltage and buck converters that output DC voltage levels that are less than the peak of the AC input supply voltage.
FIG. 1A illustrates a boost converter 100 in accordance with the prior art. Those skilled in the art will recognize that the boost converter 100 is a six-switch three-phase converter to the three input phases, .phi.A, .phi.B and .phi.C. The operation of the prior art boost converter 100 is well known, but will be discussed briefly so that the following discussion of the present invention may be more readily understood.
The input stage of the boost converter 100 comprises surge protection diodes 101-106, which clip large transient voltages on the three input lines. The diodes 101-106 do not perform any other significant function in the operation of the boost converter 100 and need not be discussed further. Current flows into and out of the boost converter 100 through inductors 111-113. Switches 121-123 and 131-133 are high speed switches that selectively connect each of the input phases to nodes N1 and N2 on either side of a capacitor 140.
The operation of the boost converter 100 may best be explained by the exemplary situation where .phi.A is the most positive voltage and XC is the most negative voltage, i.e., V.sub.A &gt;0&gt;V.sub.B &gt;V.sub.C. Current flows into the .phi.A input, through the inductor 111 and the diode 121a, and onto the capacitor 140. The circuit is completed by the rapid opening and closing (i.e., high-speed pulsing) of the switches 122, 123 at a rate of, for example, 50 kHz. Under normal operation, the voltage on the capacitor 140 is larger than the peak differences between the AC inputs.
When the switches 122, 123 are closed, the current levels in the inductors 111-113 ramp up continuously. When the switches 122, 123 are opened, the currents in the inductors 111-113 store energy on the capacitor 140, thereby maintaining the voltage level on the capacitor 140. The currents in the inductors 111-113 decrease (ramp down) while the current is being stored on the capacitor 140. The voltage on the capacitor 140 is sensed and a feedback circuit (not shown) indirectly adjusts the width of the pulses used to open and close the switches 122, 123. If the voltage level on the capacitor 140 is too low, the pulse width is increased, so that the switches 122, 123 are closed for a longer period of time. This causes the currents in the inductors 111-113 to ramp up to a higher level right before the switches 122, 123 are reopened. This higher current level stores a greater amount of charge on the capacitor 140 when the switches 122, 123 are open, thereby raising the voltage on the capacitor 140.
Conversely, if the voltage level on the capacitor 140 is too high, the pulse width is decreased, so that the switches 122, 123 are closed for a shorter period of time. This allows the currents in the inductors 111-113 to ramp up only to relatively smaller peak levels right before the switches 122, 123 are reopened. This relatively lower current level stores a smaller amount of charge on the capacitor 140 before the switches 122, 123 are reopened, thereby lowering the voltage on the capacitor 140.
When .phi.B or .phi.C becomes the highest voltage level, the operation of the boost converter 100 is virtually identical to that described above, except that different switch combinations are used to connect the three input phases to the capacitor 140.
FIGS. 1B and 1C illustrate in greater detail exemplary embodiments of unidirectional switches 121-123 and 131-133 that may be used in the prior art boost converter 100 for high power applications that: need multiple parallel switches to carry high current loads. FIG. 1B illustrates a MOSFET switch 121. FIG. 1C illustrates combined MOSFET and an insulated gate bipolar transistor (IGBT) switch 121. The switch 121 is depicted in FIGS. 1B and 1C with a diode in parallel on the right side of the switch. The parallel diode is representative of the diodes 121a-123a and 131a-133a in FIG. 1A.
The boost converter 100 has numerous shortcomings. Of the six active switches 121-123 and 131-133 in the circuit, only two operate at any one time. This is an inefficient utilization of the semiconductor devices. The six diodes must be fast recovery type diodes and each needs a snubber. The control for the switches is very complicated and interleaving pulse-width modulation operations of two or more prior art boost converters in order to reduce output ripple requires a high component count. Finally, the boost converter 100 generates a large amount of EMI noise.
Accordingly, what is needed in the art is an improved boost converter that uses its component semiconductor switches more efficiently by processing most of the principal power in a minimum number of switchers. There is a still further need in the art for a boost converter that has reduced output ripple. There is also a need in the art for a boost converter that has relatively low EMI noise, particularly with respect to equipment ground during high-speed switching.